Low-noise transmitter system and method

ABSTRACT

A low-noise transmitter architecture and method for high linearity, high output-swing systems such as Asymmetrical Digital Subscriber Line (ADSL) systems. The transmitter uses a switched-current DAC having a current source coupled to ground, followed by a resistive transimpedance amplifier (TIA). The resistance of the current source is typically large enough so that noise from an op-amp included in the TIA is not significantly amplified at the output. The current source may be passive and may include at least one resistor connected to ground. With a passive current source, portions of a signal output by the DAC enter either the current source or the resistive transimpedance amplifier, but not both, eliminating noise in the system produced by the current source.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to noise reduction in digitalcommunication systems.

2. Related Art

In many communication systems, preventing interference betweentransmitted signals and received signals is essential. An example systemis the Asymmetrical Digital Subscriber Line (ADSL) system. ADSL is ahigh-speed full duplex broadband transmission technique to connect amodem to the internet over ordinary telephone lines. For all ADSLapplications, a hybrid circuit is used to couple receive (Rx) signalsand transmit (Tx) signals from the telephone line. The hybrid circuit isalso used to separate or decouple the Tx and Rx signals from each other.

Under ideal circumstances, the hybrid circuit is able to completelydecouple the Tx and Rx signals from one another, so that the signalbeing transmitted does not interfere with the signal being received.However, due to imperfections in, for example, a telephone line andlimited performance of the hybrid circuit, the hybrid circuit will notbe able to completely decouple the Tx and Rx signals. Therefore, someamount of the residual Tx signal and Tx noise will be coupled onto theincoming Rx signal.

In order for the Rx signal to be correctly received, the interferencefrom this Tx-to-Rx coupling must be minimized. This problem isexacerbated because the Rx signal is very small when compared to the Txsignal, and any residual Tx signal or noise can corrupt the Rx signalintegrity.

As mentioned above, ADSL is only one type of communications system thatexperiences this problem. In many other communication systems, Tx noiseneeds to be minimized for similar reasons. In general, Tx noise must belimited so that the noise generated by a transmitter of a specific userdoes not interfere with the receiver of that same user (as in the caseof ADSL), another user utilizing the same communications system, oranother user using an entirely different communications system.

Although techniques exist for removing the residual Tx signal (e.g.,echo cancellation), there are no comparable techniques to remove theresidual Tx noise. Therefore, what is needed is a method and system forkeeping Tx noise at a minimum.

SUMMARY OF THE INVENTION

A low-noise transmitter for use in a communications system such as ADSLincludes a switched-current digital-to-analog converter (“DAC”) followedby a resistive transimpedance amplifier (“TIA”). The output of the DACis connected to the low-impedance input of the TIA. As a result, thereis no significant signal swing at the output of the DAC. At least onecurrent source is coupled to the DAC to establish proper common-modelevels in the transmitter. Further, noise is reduced because there is noneed to convert the DAC current into a voltage prior to feeding thesignal to the resistive TIA.

In one embodiment, the current source is active and adds current to thetransmitted signal without requiring conversion to a voltage prior toentering the TIA. In another embodiment, the current source is passive,such as a resistor connected to ground. Because the passive currentsource does not inject a signal into the system, use of a passivecurrent source further reduces noise in the system compared to theactive current source. Use of a passive current source also reducesrequired area and power consumption in the transmitter, while requiringno additional pins or external components.

In both embodiments, the value of the resistance of the current sourceis typically large enough such that the noise from an op-amp in the TIAis not significantly amplified at the output. In this manner, theoverall output noise of the transmitter is reduced compared to previoustransmitters.

Although the present invention will be described using the example of atransmitter, the present invention may also be used as an amplificationsystem in any type of digital system. Further embodiments, features, andadvantages of the present invention, as well as the structure andoperation of the various embodiments of the present invention, aredescribed in detail below with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

The accompanying drawings, which are incorporated herein and form a partof the specification, illustrate the present invention and, togetherwith the description, further serve to explain the principles of theinvention and to enable a person skilled in the pertinent art to makeand use the invention.

FIG. 1 is a diagram of an example transmitter that uses aswitched-capacitor digital-to-analog converter (“DAC”) and a resistivegain amplifier (“RGA”).

FIG. 2 is a diagram of another example transmitter that uses an RGA.

FIG. 3 is a diagram of an example transmitter that uses aswitched-current DAC and an RGA.

FIG. 4 is a diagram of an example transmitter that uses aswitched-current DAC with an active current source and a resistivetransimpedance amplifier (“TIA”).

FIG. 5 is a diagram of an example transmitter that uses aswitched-current DAC with a passive current source and a resistive TIA.

The present invention will be described with reference to theaccompanying drawings. The drawing in which an element first appears istypically indicated by the leftmost digit(s) in the correspondingreference number.

DETAILED DESCRIPTION OF THE INVENTION

While specific configurations and arrangements are discussed, it shouldbe understood that this is done for illustrative purposes only. A personskilled in the pertinent art will recognize that other configurationsand arrangements can be used without departing from the spirit and scopeof the present invention. It will be apparent to a person skilled in thepertinent art that this invention can also be employed in a variety ofother applications. For example, one of ordinary skill in the relevantart will recognize that the amplification circuit described herein neednot be used solely as an ADSL transmitter, but that it may be utilizedas a low-noise amplifier in any application where an analog signal isamplified from either a digital or analog source.

When used as an ADSL transmitter, the specifications of the ADSL systemrequire the transmitter to be able to drive a large signal with alinearity exceeding 90 dB. In order to meet this high-swing,high-linearity requirement, a digital-to-analog converter (“DAC”)followed by one or more stages of gain is used in various ADSLtransmitter architectures.

Switched-Capacitor DAC Followed by Resistive Gain Amplifier

FIG. 1 is a diagram of an example transmitter 100. Transmitter 100includes a switched-capacitor DAC (“SC-DAC”) 102 and a resistive gainamplifier (“RGA”) 104. RGA 104 includes an op-amp 106, resistors 108 and110, and feedback resistors 112 and 114. Resistors 108 and 110 each havea resistance R₁, while feedback resistors 112 and 114 each have aresistance R₂. The gain of RGA 104 is based on a ratio of feedbackresistors 112 and 114 to resistors 108 and 110.

SC-DAC 102 outputs a voltage-based signal V_(DAC). Op-amp 106 amplifiesvoltage-based signal V_(DAC), the gain set by the ratio of feedbackresistors 112 and 114 and resistors 108 and 110, and then outputs avoltage-based output signal V_(OUT).

SC-DACs, such as SC-DAC 102, have relatively high noise levels due tothe sampled-time nature of the switched capacitors. It is well knownthat switched-capacitor circuits have kT/C noise, which is noise due tofluctuation of charge stored on capacitance C. In order to reduce thepower spectral density of this kT/C noise, either large capacitors mustbe used or very high oversampling ratios must be used. Large capacitorsare impractical for integrated circuits because they require a largearea. The oversampling ratio for an SC-DAC is limited by the speed ofthe circuit technology. Increasing the oversampling ratio of an SC-DACusually requires a significant increase in power consumption.

Additionally, RGAs, such as RGA 104, are typically power-consuming andnoisy devices. Since SC-DAC 102 typically has relatively low outputswing, RGA 104 must have a reasonable gain. As shown in the followingequations, the noise at the output of RGA 104 contributed by op-amp 106is amplified by RGA 104 in a similar manner to an input signal V_(DAC):$\begin{matrix}{\frac{\overset{\_}{V_{OUT}^{2}}}{\Delta\quad f} = {2 \cdot \left( {{4\quad{{kT} \cdot R_{1} \cdot A_{RGA}^{2}}} + {4{{kT} \cdot R_{2}}} + {\frac{\overset{\_}{V_{N}^{2}}}{\Delta\quad f} \cdot \left( {1 + A_{RGA}} \right)^{2}} + {\frac{\overset{\_}{V_{DAC}^{2}}}{\Delta\quad f} \cdot A_{RGA}^{2}}} \right)}} & \left( {{Eq}.\quad 1} \right) \\{where} & \quad \\{A_{RGA} = {\frac{V_{OUT}}{V_{DAC}} = {\frac{R_{2}}{R_{1}}.}}} & \left( {{Eq}.\quad 2} \right)\end{matrix}$

A_(RGA) is the amplitude of the amplifier noise, and is a function ofthe gain of the amplifier. V_(N) is the voltage input into op-amp 106. kis Boltzmann's constant, T is the temperature in °K, and C is thecapacitance of the SC-DAC. Each of the resistor sets (i.e., resistors108 and 110, and feedback resistors 112 and 114) adds noise to thesystem, as shown by the inclusion of R₁ and R₂ in Eq. 1. This puts astringent noise specification upon op-amp 106 within RGA 104.

FIG. 2 is a diagram of an example transmitter 200. Transmitter 200 issimilar in architecture to transmitter 100, but uses a genericlow-impedance input source 202 in place of an SC-DAC. Transmitter 200also includes an RGA 204 having an op-amp 206, resistors 208 and 210,feedback resistors 212 and 214, and voltage sources 216 and 218.Resistors 208 and 210 are approximately equal, each having a resistanceR₁. Feedback resistors 212 and 214 are also approximately equal to eachother, each having a resistance R₂.

Op-amp 206 outputs an amplified signal V_(OUT), wherein the gain isbased on the ratio of feedback resistors 212 and 214 to resistors 208and 210.

Switched-Current DAC with Resistive Load followed by RGA

FIG. 3 is a diagram of an example transmitter 300 that uses aswitched-current DAC (“SI-DAC”) 302 with an RGA 304 and a resistive loadcreated by resistors 316 and 318. Like RGA 104 and RGA 204, RGA 304includes an op-amp 306, resistors 308 and 310, and feedback resistors312 and 314. Since an SI-DAC is used in place of an SC-DAC, resistors316 and 318 are used to set the voltage swing at the output of theSI-DAC. Compared to the example of FIGS. 1 and 2, SI-DAC 302 followed byRGA 304 has a lower noise. This is because SI-DAC 302 does not add thekT/C noise of an SC-DAC. Even so, this architecture has severallimitations.

First, transmitter 300 utilizes an RGA. Therefore, all the problemsdescribed with respect to RGA 104 and RGA 204 apply to RGA 304.Specifically, the signal is still affected by two sets of resistors(i.e., resistors 308 and 310, and feedback resistors 312 and 314) tocreate an output signal, which adds noise to the system.

Also, a limitation with high-linearity SI-DACs is that their outputswing is typically severely limited. As shown in Eq. 1, if V_(OUT) isassumed to be fixed, requirements are imposed on the product of V_(DAC)and A_(RGA). A large A_(RGA) increases the total output noise of thetransmitter. Therefore, transmitter 300 requires a reasonably largeV_(DAC). This can be difficult to achieve while maintaining highlinearity.

In each of transmitters 100, 200, and 300, the act of converting thesignal between current and voltage domains results in a large amount ofoutput noise due to the resistors. Also, resistance R₁ from theresistors presents a load that needs to be driven by the previous stage.Thus, if the resistors could be removed, the noise and total powerconsumption of the overall circuit would be significantly reduced.

Switched-Current DAC with Active Current Source

FIG. 4 is a diagram of an example transmitter 400 according to anembodiment of the present invention. Transmitter 400 includes an SI-DAC402, active current sources 404 and 406, and a resistive transimpedanceamplifier (“TIA”) 408. TIA 408 includes an op-amp 410, feedbackresistors 412 and 414, and voltage sources 416 and 418. SI-DAC 402converts an input digital signal to an analog current-based signal.Because a TIA can convert a current-based signal to a voltage-basedsignal within the TIA itself, there is no need for resistors to performthis step before the signal is input to the TIA. The current-basedsignal from SI-DAC 402 can be input directly into resistive TIA 408.

Thus, an SI-DAC with active current sources 404 and 406 followed by aresistive TIA avoids many of the problems described above. For example,an input signal in transmitter 400 is affected by one less resistor set.This removes any noise associated with resistors in the examples ofFIGS. 1, 2, and 3. Instead of amplifying an input voltage, resistive TIA408 acts as a current-to-voltage converter and converts thecurrent-based signal from SI-DAC 402 into a voltage-based signal.Resistive TIA 408 also amplifies the signal to produce an output signalV_(OUT). Thus, as shown in the following noise equation, theoutput-referred amplifier noise is no longer a function of the gain ofthe amplifier stage: $\begin{matrix}{\frac{\overset{\_}{V_{OUT}^{2}}}{\Delta\quad f} = {2 \cdot {\left( {{4\quad{{kT} \cdot R_{TIA}}} + \frac{\overset{\_}{V_{N}^{2}}}{\Delta\quad f} + {\frac{\overset{\_}{I_{CS}^{2}}}{\Delta\quad f} \cdot R_{TIA}^{2}} + {\frac{\overset{\_}{I_{DAC}^{2}}}{\Delta\quad f} \cdot R_{TIA}^{2}}} \right).}}} & \left( {{Eq}.\quad 3} \right)\end{matrix}$

Further, the transimpedance gain is equal to the resistance of feedbackresistors 412 and 414, or R_(TIA), rather than a ratio of feedbackresistors to resistors.

The inputs to the TIA create a virtual ground node at node 420. SinceSI-DAC 402 is feeding into a virtual ground node, the signal swing atthe output of SI-DAC 402 is kept small, even if a large output swingexists. Therefore, high linearity of the transmitter output V_(OUT) canbe maintained. Current sources 404 and 406 shown in FIG. 4 maintainproper common-mode levels throughout the transmitter and drive theanalog current-based signal.

Current sources 404 and 406 shown in the embodiment of FIG. 4 arecreated from active devices. Because the active devices inject a signalinto a node 420 between SI-DAC 402 and resistive TIA 408, they maycontribute noise to the output of transmitter 400. In addition, theactive devices may be biased by a network of similar active devices.These bias devices could contribute additional noise to the output. Thenoise produced by these devices can cause difficulties at lowfrequencies, where the flicker noise of the active devices can dominate,since the I_(cs) ²/Δf term becomes very large. Since the ADSL transmitband, for example, is at approximately 100 kHz, the flicker noise of theactive devices may overpower the Tx signal in such an application.

In order to limit the total noise contributed by the active devices,several steps can be taken. Flicker noise can be reduced by increasingthe area of the active devices, and/or the noise from the bias devicescan be effectively eliminated by attaching a large bypass capacitor tothe bias network. However, increased area translates into increasedparasitic capacitances. Further, attaching a bypass capacitor wouldrequire an additional pin on the package and an additional discretecomponent. Both of these steps would incur additional system costs.

Switched-Current DAC with Passive Current Source

FIG. 5 is a diagram of an example transmitter 500 according to anotherembodiment of the present invention. Transmitter 500 includes an SI-DAC502 with passive current sources 504 and 506 followed by a resistive TIA508. TIA 508 includes an op-amp 510, feedback resistors 512 and 514, andvoltage sources 516 and 518. Passive current sources 504 and 506 may be,for example, resistors connected to ground which are used to establishproper common-mode levels. In the example of FIG. 5, each of passivecurrent sources 504 and 506 has a resistance R_(CS).

Compared to transmitter 400, transmitter 500 has reduced noise due tothe substitution of passive current sources for active current sources.Like transmitter 400, the signal in transmitter 500 is processed by oneless resistor set compared to transmitters 100, 200, and 300, due to thelack of resistors. As shown in FIG. 5, current sources 504 and 506branch off of a signal path 520 between SI-DAC 502 and TIA 508. Althougha portion of the signal output by SI-DAC 502 may enter passive currentsources 504 and 506, that portion of the signal is common-mode and hencenot processed by TIA 508. Therefore, the differential transfer functionof the signal is unaffected by the presence of passive current sources504 and 506.

Thus, the total output noise of the embodiment of FIG. 5 is:$\begin{matrix}{\frac{\overset{\_}{V_{OUT}^{2}}}{\Delta\quad f} = {2 \cdot \left( {{4\quad{{kT} \cdot R_{TIA}}} + {\frac{\overset{\_}{V_{N}^{2}}}{\Delta\quad f} \cdot \left( {1 + \frac{R_{TIA}}{R_{CS}}} \right)^{2}} + {\frac{4{kT}}{R_{CS}} \cdot R_{TIA}^{2}} + {\frac{\overset{\_}{I_{DAC}^{2}}}{\Delta\quad f} \cdot R_{TIA}^{2}}} \right)}} & \left( {{Eq}.\quad 4} \right) \\{where} & \quad \\{\frac{R_{TIA}}{R_{CS}} < 1.} & \left( {{Eq}.\quad 5} \right)\end{matrix}$

The value of resistance R_(CS) used in passive current sources 504 and506 is typically large enough such that noise from op-amp 510 in TIA 508is not significantly amplified at the output. For ADSL applications,R_(CS) may be, for example, 1 kΩ. Also, passive current sources 504 and506 have negligible flicker noise. Therefore, at low frequencies, theoverall output noise of this topology is typically far lower than insystems described above. This allows improved application at lowerfrequencies than with previous systems. For instance, for ADSLapplications the frequency range may be, for example, 20 kHz to 276 kHz.However, one of ordinary skill in the relevant art(s) will recognizethat op-amp 510 may be tuned to any frequency for the same or alternateapplications without departing from the spirit and scope of the presentinvention. Also, passive current sources 504 are much smaller in areaand power consumption than active current sources, and require noadditional pins or external components. Therefore, the embodiment ofFIG. 5 is amenable to a low-cost, low-noise system, and effectivelyreduces noise from a Tx signal.

The embodiments described with reference to FIGS. 4 and 5 are alsodifferent from that of transmitter 300 in that the performance of theembodiments is quite different. First, transmitters 400 and 500 injectthe signal from the SI-DAC into the TIA virtual ground formed by thecurrent sources, so there is no significant signal swing at the outputof the SI-DAC. Additionally, a resistive TIA amplifies op-amp noise farless than, for example, RGA 304 in transmitter 300. In transmitter 500,the purpose of resistors R_(cs) used as passive current sources 504 and506 is not to convert the current from SI-DAC 502 into a voltage;instead, resistors R_(cs) are used as passive current sources toestablish proper common mode. In such ways, the Tx noise in each ofexample transmitters 400 and 500 is reduced compared to exampletransmitters 100, 200, and 300.

Conclusion

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample only, and not limitation. It will be apparent to persons skilledin the relevant art that various changes in form and detail can be madetherein without departing from the spirit and scope of the invention.Thus, the breadth and scope of the present invention should not belimited by any of the above-described exemplary embodiments, but shouldbe defined only in accordance with the following claims and theirequivalents.

1. A transmitter, comprising: (a) a digital-to-analog converter (DAC);(b) a current source coupled to an output of the DAC and to ground; and(c) a resistive transimpedance amplifier coupled to the output of theDAC, wherein a gain of the resistive transimpedance amplifier isapproximately equal to a feedback resistance within the resistivetransimpedance amplifier.
 2. The transmitter of claim 1, wherein thecurrent source is an active current source.
 3. The transmitter of claim1, wherein the current source is a passive current source.
 4. Thetransmitter of claim 3, wherein the current source comprises at leastone resistor coupled to ground.
 5. The transmitter of claim 4, wherein aresistance of the current source is determined by the amount of noiseproduced by the operational amplifier.
 6. The transmitter of claim 5,wherein the resistance of the current source is approximately 1 kΩ. 7.The transmitter of claim 1, wherein the current source establishes acommon mode level for the transmitter.
 8. The transmitter of claim 1,wherein the resistive transimpedance amplifier comprises: (i) anoperational amplifier; (ii) a first feedback resistor coupled inparallel to a positive path of the operational amplifier; and (iii) asecond feedback resistor coupled in parallel to a negative path of theoperational amplifier.
 9. The transmitter of claim 8, wherein the firstand second feedback resistors are variable resistors.
 10. Anamplification system for amplifying a digital signal, comprising: (a) adigital-to-analog converter (DAC); (b) a current source coupled to theoutput of the DAC and to ground; and (c) a resistive transimpedanceamplifier coupled to the output of the DAC, wherein a gain of theamplification system is approximately equal to a feedback resistancewithin the resistive transimpedance amplifier.
 11. The amplificationsystem of claim 10, wherein the digital signal is an ADSL signal. 12.The amplification system of claim 10, wherein the current source is anactive current source.
 13. The amplification system of claim 10, whereinthe current source is a passive current source.
 14. The amplificationsystem of claim 13, wherein the current source comprises at least oneresistor coupled to ground.
 15. The amplification system of claim 14,wherein a resistance of the current source is determined by the amountof noise produced by the operational amplifier.
 16. The amplificationsystem of claim 15, wherein a resistance of the current source isapproximately 1 kΩ.
 17. The amplification system of claim 10, whereinthe resistive transimpedance amplifier comprises: (i) an operationalamplifier; (ii) a first feedback resistor coupled in parallel to apositive path of the operational amplifier; and (iii) a second feedbackresistor coupled in parallel to a negative path of the operationalamplifier.
 18. A transmitter, comprising: (a) a digital-to-analogconverter (DAC); (b) a resistive transimpedance amplifier; (c) a signalpath connecting the DAC and the resistive transimpedance amplifier; and(d) a passive current source coupled between the signal path and ground,wherein portions of a signal traversing the signal path enter either thecurrent source or the resistive transimpedance amplifier.
 19. Thetransmitter of claim 18, wherein the current source comprises at leastone resistor coupled to ground.
 20. The transmitter of claim 19, whereina resistance of the current source is determined by the amount of noiseproduced by the operational amplifier.
 21. The transmitter of claim 20,wherein a resistance of the current source is approximately 1 kΩ. 22.The transmitter of claim 18, wherein the resistive transimpedanceamplifier comprises: (i) an operational amplifier; (ii) a first feedbackresistor coupled in parallel to a positive path of the operationalamplifier; and (iii) a second feedback resistor coupled in parallel to anegative path of the operational amplifier.
 23. The transmitter of claim22, wherein a gain of the transmitter is approximately equal to afeedback resistance within the resistive transimpedance amplifier.
 24. Amethod of amplifying a digital signal, comprising: (a) converting thedigital signal to an analog voltage-based signal; (b) amplifying theanalog voltage-based signal using a resistive transimpedance amplifierwhose gain is approximately equal to a feedback resistance within thetransimpedance amplifier; and (c) outputting the amplified voltage-basedsignal.
 25. The method of claim 24, further comprising: (d) driving theanalog voltage-based signal with a current source connected to ground.26. The method of claim 25, wherein said step (d) comprises: driving theanalog voltage-based signal with an active current source.
 27. Themethod of claim 25, wherein said step (d) comprises: driving the analogvoltage-based signal with a passive current source.